The present invention relates in general to electromagnetic test systems. More particularly, the present invention relates to antenna test range design. Still more particularly, the present invention relates to the creation of so called farfield test conditions in a very limited space, as for example in anechoic chambers.
The principles of antenna range design are discussed in detail by Hollis et al, "Microwave Antenna Measurements" (Scientific-Atlanta, Inc., 1969), but the following summary of portions thereof will assist in understanding the invention. When testing any radiating devices or device/systems receiving electromagnetic energy, the ideal test environment for determining far-field performance is to provide a plane wave of uniform amplitude and phase to illuminate the test aperture. Various approaches to simulation of this ideal electromagnetic environment have led to the evolution of two basic types of electromagnetic test facilities:
(1) Free-space Ranges PA1 (2) Reflection Ranges
Free-space ranges are those in which an attempt is made to suppress or remove the effects of all surroundings, including the range surface or surfaces, on the wavefront which illuminates the test antenna. This suppression is sought through one or more of such factors as (a) directivity and sidelobe suppression of the source antenna and test antenna, (b) clearance of the line of sight from the range surface, (c) redirection or absorption of energy reaching the range surface, and (d) special signal processing techniques such as tagging by modulation of the desired signal or by use of short pulses.
The typical geometries associated with the free-space approach include the elevated range, the slant range, the rectangular anechoic chamber, and, above certain limiting frequencies, the tapered anechoic chamber. A recent development in this area is the compact range, in which the test antenna is illuminated by collimated energy in the aperture of a larger point--or line--focus antenna.
Reflection ranges are designed to make use of energy which is reradiated from the range surface(s) to create constructive interference with the direct-path signal in the region about the test aperture. The geometry is controlled so that a small, essentially symmetric amplitude taper is produced in the illuminating field. The two major types of reflection ranges in use are the ground reflection range and, for low frequencies, the tapered anechoic chamber.
For either basic type of range, the fundamental electromagnetic design criteria deal with control of five factors: (A) Inductive or radiation coupling between antennas; (B) Phase curvature of the illuminating wavefront; (C) Amplitude taper of the illuminating wavefront; (D) Spatially periodic variations in the illuminating wavefront caused by reflections; and (E) Interference from spurious radiating sources.
Items A through D primarily establish the dimensional requirements in the range design, and limiting values of source-antenna directivity. Item E must be considered in the overall design.
At lower microwave frequencies, effects of inductive coupling between the source antenna and the test antenna must be considered. Such effects are usually considered negligible when the criterion EQU R.ltoreq.10.lambda. (1)
is satisfied, where R is the separation between antennas and .lambda. is the wavelength. The criterion is based on the field equations for an elemental electric dipole, from which the ratio of the amplitude of the induction field to that of the radiation field is seen to be EQU .rho..epsilon.=(.lambda./2.pi.R) (2)
At R.ltoreq.10.lambda., .rho..epsilon..ltoreq.1/20.pi., and the criterion is seen to be equivalent to the requirement that EQU 20 log (.rho..epsilon.).ltoreq.-36 decibels (3)
The effect of curvature of the incident phase front is a most important one. The principal difficulty is that the generally accepted criteria is that the minimum range length acceptable is determined by the relationship EQU R.ltoreq.(2D.sup.2 /.lambda.) (4)
Where D is the diameter or maximum dimension of the test item (e.g. aperture).
For apertures in excess of twelve (12) inches at X-Band or above (&gt;8 GHz) the range length requirement is longer than sixteen (16) feet, which is the maximum length of the most common size rectangular chambers used for measuring low gain antennas. Due to the high cost of absorbing materials, larger chambers are probibitively expensive, and outdoor ranges are not always conveniently available, either due to lack of space or weather.
More particularly, in the absence of reflections, the phase variation of the field over the aperture of a receiving antenna of a given size and operating at a given frequency depends almost entirely on the separation between the source antenna and the receiving antenna. If the receiving antenna is in the far zone of the transmitting antenna, the phase front of the approaching wave deviates very little from a section of a sphere centered on the transmitting antenna over the major portion of the main lobe.
It can be shown that over a planar receiving aperture the variation of the phase of the incident field is caused almost entirely by deviation of the test aperture from the sphere on the transmitting antenna if the receiving antenna subtends less than a half-power beamwidth of the transmitting antenna's wave front. In practice, the antenna under test will subtend considerably less than a half-power beamwidth in order to reduce error from mutual coupling and from amplitude taper of the incident field over the test aperture.
An expression for the phase deviation over a planar test aperture can be determined from FIG. 1. Since EQU R.sup.2 +(D.sup.2 /4)=(R+.DELTA.R).sup.2 ( 5) EQU .DELTA.R.apprch.(D.sup.2 /8R) (6)
if R.sup.2 is neglected. The corresponding phase deviation is given by EQU .DELTA..phi.=(2.pi..DELTA.R/.lambda.)=(.pi.D/4.lambda.R) radians (7)
A commonly employed criterion for determining the minimum allowable separation between the source antenna and the antenna under test is to restrict .DELTA..phi. to a maximum of .pi./8 radian, or 22.5 degrees. Under this condition, R.ltoreq.2D.sup.2 /.lambda. (Eq.4), there will be a significant departure of the nulls of the radiation pattern and the location and levels of the minor lobes from their infinite-range values. The amount of the deviation depends on the original side-lobe level and structure. Calculations have shown that for a range of 2D.sup.2 /.lambda., the first null of the pattern produced by a rectangular aperture with uniform illumination has relative level of about -23 decibels instead of -.infin. decibels. This theoretical deviation is due solely to phase-error effects; the incident-wave amplitude over the test aperture was assumed constant. The infinite range pattern in the above case has a (sin x/x ) configuration, with a relative first-lobe level of about -13 decibels.
The effect of amplitude taper over the test aperture must also be considered. For accuracy in simulated far zone measurements, the illuminating field must be sufficiently constant in amplitude both along the line-of-sight and in planes normal to the line-of-sight.
Consider an antenna under test on receiving, which has a maximum dimension, L, of its active region along the line-of-sight. If the separation between the source of antenna and the center of the active region is R.sub.o, then the ratio p.rho. of the power density at the forward extreme of the active region to that at the rear is given by ##EQU1## Severe axial variations of the illuminating field can cause measurement error, particularly in the minor lobe structure of radiation patterns. For most antenna types which have significant depth to their active regions, such error is usually negligible when the power density over the region is constant to within one decibel. This condition corresponds to an approximate restraint EQU R.sub.o .ltoreq.10L (9)
The criterion for such structures as high-gain disc-on-rod antennas, often is more restrictive than the greater of the previously discussed range-length criteria which were based on suppression of inductive coupling and phase curvature.
The effect of amplitude taper of the incident field over a plane normal to the line of sight and adjacent to the test aperture can be considered from the viewpoint of reciprocity. Variation of the amplitude of the field over the aperture on receiving is analogous--within the accuracy of the aperture field approach--to modification of the aperture illumination by the primary feed on transmitting. For example, the pattern of an antenna whose feed would produce an aperture illumination f(.theta.,r) on transmitting, where (.theta.,r ) indicates position in the aperture, if illuminated on receiving by a source antenna which produces over the test aperture an amplitude taper g(.theta.,r), the measured pattern would be analogous to that of a transmitting antenna illuminated by a feed which produces an illumination of f(.theta.,r)g(.theta.,r) over the aperture. If g(.theta.,r) is constant in amplitude and phase over the aperture, the measured pattern will be the same as the infinite-range pattern for the illuminations f(.theta.,r). The greater g(.theta.,r) deviates from constant, the greater will be the deviation of the measured pattern from the infinite-range pattern. The quantitative effect of nearly constant functions g(.theta.,r) cannot be determined, however, without assumption of f(.theta.,r).
Calculations indicate that for a 0.5 dB amplitude taper across the unit under test a 0.15 dB decrease in measured gain will result. If the taper is equal to 0.25 dB, the error is less than 0.1 dB.
If a source antenna is employed which is calculated to produce a taper of the field over the test aperture, it is essential that the transmitting antenna be directed such that the peak of its beam is centered on the antenna under test to prevent excessive and asymmetrical illumination taper with a resultant increase in the measuring error. It is important to note that error from symmetrical amplitude taper within the accepted criterion of 0.25 decibel does not produce a defocusing type of error, but rather a small modification of the measured side-lobe levels and an error in measured gain.
As discussed above, the testing of microwave antennas usually requires that the device under test be illuminated by a uniform plane electromagnetic wave. However, the creation of such a wave can be a difficult task. Conventional techniques require that a transmitting antenna be located at a sufficient distance from the test antenna such that its spherical wavefront closely approximates a uniform plane wave incident upon the test device.
Since ranges of several hundred to several thousand feet are often required to satisfy the dD.sup.2 /.lambda.criteria, far-zone measurements usually are taken on outdoor installations which are subject to adverse weather conditions and changing range effects. Further, the cost of land for such ranges (adjacent or at lease near the manufacturing facility) can be very high, and security can be a problem. Small antenna or targets may be tested adequately in anechoic chambers, but since large antennas (large in terms of wavelength) require long ranges, the cost of a chamber for such tests becomes prohibitively high.
The technique described in U.S. Pat. No. 3,302,205 enables measurements with full-size antennas or fair size targets to be made on indoor `compact ranges`, and is currently available. A range reflector and a special feed system close to the test device are used to produce incident plane waves and far-zone results are obtained. A properly focused parabolic-type reflector collimates the rays and thus produces a plane wave across its aperture. This wave is not uniform due to the illumination taper of the feed horn, and due to space-attenuation effects. However, a properly selected feed will generate a wave which is approximately uniform over an acceptable area. It is this area of an approximately uniform plane wave that is used on compact ranges to illuminate the antenna under test. This patent also speaks of prior efforts to produce plane wave illumination with lenses, noting that unsatisfactory results were obtained due to amplitude distortion caused by random and uncontrolled reflections.
Dielectric lenses and cones are known per se, and have been used in microwave and other transmission systems for many years. The principles of design of a dielectric lens were described by Silver, "Microwave Antenna Theory and Design (McGraw Hill, 1949)." The use of conical dielectric horns to improve the efficiency of microwave reflector and horn antennas is disclosed in U.S. Pat. No. 3,430,244, No. 3,414,903 and No. 6,611,391 of Bartlett et al. In the U.S. Pat. No. 3,414,903 patent, a cone of low dielectric constant material is used in conjunction with a lens of higher dielectric constant material to produce a transmitter with a substantial side-lobe depression and a planar wave front.
Understanding of the invention will be facilitated by considering the following analysis of two simple lens types: the planoconvex lens and the meniscus lens. The profiles of these lenses are known and may be found in Silver (op. cit.)
For a planoconvex lens (from FIG. 2): ##EQU2##
where n=.sqroot..epsilon./.epsilon.o=relative permittivity of the lens, also defined as the refractive index or dielectric constant, and f=focal length of the lens.
For a meniscus lens (see FIG. 3): ##EQU3##
The focal length of the lens is the distance from the phase center of the illuminating antenna to the center of the closest point on the planoconvex lens and the center of the lens farthest from the phase center of the meniscus lens.
To generate the lens geometry given the focal length and the relative permittivity, the equations are rearranged as follows:
Planoconvex Lens: ##EQU4##
Meniscus Lens: ##EQU5##
To be useful, the lens of the invention must not degrade the performance of the test region consistent with the following guide lines:
The lens operates only on the propagation constant in the wave equation. That is, the wavefront is delayed more at the center of the lens than at the edges. Thus, a spherical wavefront striking the lens delays the center of the wavefront greater in the center, thus causing the emerging wavefront to be all in line or uniform as it is called in the antenna testing field. This approximate uniform field is then used to illuminate a test device which is typically an antenna. The result is that the test antenna "sees" far field test conditions (uniform amplitude and phase) and thus behaves as if it were one. Thus, by inserting a lens between a source antenna and a test antenna the distance can be shortened because the lens provides the delay needed to cause a planar wavefront to be created.
When the wave passes through the lens it encounters the boundaries between the lens and free space. Due to the difference in the dielectric constant of air and the lens, a portion of the wave is reflected at each boundary. These do not occur uniformly, since the lens geometry varies in thickness; thus the uniformity of the wave actually reaching the test region is distorted by the reflections and refractions at the lens boundaries. This distortion appears as ripples on the amplitude properties of the wave in the test region. If the dielectric constant is low enough, however, then this ripple is well within acceptable values.
Unfortunately, solid dielectrics traditionally used in such applications (ethyl cellulose, polyethylene, polystyrene, polyisobutylene or methyl methacrylate) all have a dielectric constant that is much too high to be of use. It is believed that efforts of prior workers to create plane wave conditions with lenses could have been unsatisfactory because of a failure to recognize (1) the need to use a lens of very low dielectric constant and (2) the need to independently minimize amplitude distortion (e.g., reflections) by using absorbent materials and proper selection of the source antenna in the test range.
In addition to boundary reflections referred to supra, the test antenna will reflect a certain amount of energy back to the lens, which re-reflects it back to the antenna, as an extraneous signal of arbitrary phase. This can cause serious measurement problems. More particularly, no antenna or other radiating device is 100% efficient; most will capture 50-70% of the incident energy, and some are only 25% efficient. The energy not captured is scattered from the surface of the structure, with much of it travelling back toward the transmitting antenna. Indeed, when carrying out radar cross-section measurements, all of the energy is reflected, since the target acts as short-circuited antenna. In a conventional free space test range, the reflected energy is attenuated in space and does not get back to the test device. With a lens test system, in accordance with the present invention, a reflection will occur. More particularly, the higher the dielectric constant of the lens, the more compact the test range may be, but the measurement errors caused by re-reflection will become more serious.